Successive approximation sigma delta analog-to-digital converters

ABSTRACT

An A/D converter including first and second A/D converters and a recombination module. The first A/D converter receives an analog input signal, converts the analog input signal to a first digital signal, and includes a successive approximation module, which performs a successive approximation to generate the first digital signal. The second A/D converter converts an analog output of the first A/D converter to a second digital signal. The analog output of the first A/D converter is generated based on the analog input signal. The second A/D converter is a fine conversion A/D converter relative to the first A/D converter. The second A/D converter performs the delta-sigma conversion process and includes a decimation filter that suppresses noise which reduces amplification and power consumption requirements of the first A/D converter and performs a delta-sigma decimation process to generate the second digital signal based on the analog output of the first A/D converter.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims is a continuation of U.S. Non-Provisionalapplication Ser. No. 15/204,365, filed Jul. 7, 2016, which claims thebenefit of U.S. Provisional Application No. 62/189,872, filed Jul. 8,2015, U.S. Provisional Application No. 62/234,148, filed Sep. 29, 2015and U.S. Provisional Application No. 62/200,823, filed Aug. 4, 2015. Theentire disclosures of the applications referenced above are incorporatedherein by reference.

FIELD

The present disclosure relates to topologies and methods for reducingthe size and/or for reducing power demands of analog-to-digitalconverters.

BACKGROUND

A successive approximation register (SAR) analog-to-digital converter(ADC) converts an analog waveform into a discrete digital representationvia a binary search through all possible quantization levels beforefinally converging upon a digital output for each conversion. A SAR ADCperforms a successive approximation algorithm (or sometimes referred toas “a binary search algorithm”) to provide a binary code. When theapproximation is completed, the SAR ADC outputs an estimated digitaloutput indicating the binary code.

SUMMARY

A hybrid digital-to-analog converter is provided including a firstdigital-to-analog converter and a second digital-to-analog converter.The first digital-to-analog converter is configured to (i) receive adigital input signal having an input voltage, and (ii) convert a firstmost-significant-bit of multiple bits of the digital input signal to beconverted to an analog signal. The first digital-to-analog converterincludes first capacitors. The first capacitors are charged by the inputvoltage and reference voltages during a sampling phase of the digitalinput signal. Charges of the first capacitors are shared duringsuccessive approximations of a first one or more bits of the digitalinput signal received by the hybrid digital-to-analog converter toprovide the analog signal. The second digital-to-analog converter isconfigured to convert a first least-significant-bit of the bits of thedigital input signal to be converted to the analog signal. The seconddigital-to-analog converter includes second capacitors. The secondcapacitors are charged based on a common mode voltage during thesampling phase of the digital input signal. The second digital-to-analogconverter is to perform charge redistribution by connecting the secondcapacitors to receive the reference voltages during successiveapproximations of a second one or more bits of the digital input signal.

In other features, a method is provided and includes: receiving adigital input signal having an input voltage at a firstdigital-to-analog converter; converting a first most-significant-bit ofmultiple bits of the digital input signal to be converted to an analogsignal via the a first digital-to-analog converter; and charging firstcapacitors of the a first digital-to-analog converter by the inputvoltage and reference voltages during a sampling phase of the digitalinput signal. The method further includes: sharing charges of the firstcapacitors during successive approximations of a first one or more bitsof the digital input signal received by the hybrid digital-to-analogconverter to provide the analog signal; converting a firstleast-significant-bit of the bits of the digital input signal to beconverted to the analog signal at a second digital-to-analog converter;charging second capacitors of the second digital-to-analog converterbased on a common mode voltage during the sampling phase of the digitalinput signal; and performing charge redistribution via the seconddigital-to-analog converter by connecting the second capacitors toreceive the reference voltages during successive approximations of asecond one or more bits of the digital signal.

In other features, an analog-to-digital converter is provided andincludes a hybrid digital-to-analog converter, an amplifier, a latch anda successive approximation module. The hybrid digital-to-analogconverter includes a first digital-to-analog converter and a seconddigital-to-analog converter. The first digital-to-analog converter isconfigured to (i) receive a digital input signal having an inputvoltage, and (ii) convert a first most-significant-bit of multiple bitsof the digital input signal to be converted to an analog signal, wherethe first digital-to-analog converter comprises first capacitors. Thefirst capacitors are charged by the input voltage and reference voltagesduring a sampling phase of the digital input signal. Charges of thefirst capacitors are shared during successive approximations of a firstone or more bits of a digital signal received by the hybriddigital-to-analog converter to provide the analog signal. The seconddigital-to-analog converter is configured to convert a firstleast-significant-bit of the bits of the digital input signal to beconverted to the analog signal. The second digital-to-analog converterincludes second capacitors. The second capacitors are charged based on acommon mode voltage during the sampling phase of the digital inputsignal. The second digital-to-analog converter is to perform chargeredistribution by connecting the second capacitors to receive thereference voltages during successive approximations of a second one ormore bits of the digital signal. The amplifier or integrator isconfigured to amplify or integrate the analog signal. The latch isconfigured to latch an output of the amplifier or integrator. Thesuccessive approximation module is configured to (i) receive an outputof the latch, and (ii) perform the successive approximations of thefirst one or more bits of the digital signal and the successiveapproximations of the second one or more bits of the digital signal.

In other features, an analog-to-digital converter is provided andincludes a first analog-to-digital converter, a second analog-to-digitalconverter and a combination module. The first analog-to-digitalconverter is configured to receive an analog input signal and convertthe analog input signal to a first digital signal. The firstanalog-to-digital converter includes a successive approximation module.The successive approximation module is configured to perform asuccessive approximation to generate the first digital signal. Thesecond analog-to-digital converter is configured to convert an analogoutput of the first analog-to-digital converter to a second digitalsignal. The analog output of the first analog-to-digital converter isgenerated based on the analog input signal. The second analog-to-digitalconverter is a fine conversion analog-to-digital converter relative tothe first analog-to-digital converter. The second analog-to-digitalconverter comprises a decimation filter. The decimation filter isconfigured to: suppress noise which reduces amplification and powerconsumption requirements of the first digital-to-analog converter; andperform a delta-sigma decimation process to generate the second digitalsignal based on the analog output of the first analog-to-digitalconverter. The combination module is configured to combine the firstdigital signal and the second digital signal to provide a resultantoutput signal.

In other features, a method is provided and includes: receiving ananalog input signal and converting the analog input signal to a firstdigital signal at a first analog-to-digital converter; performing asuccessive approximation to generate the first digital signal via thefirst analog-to-digital converter; and converting an analog output ofthe first analog-to-digital converter to a second digital signal via asecond analog-to-digital converter, where the second analog-to-digitalconverter is a fine conversion analog-to-digital converter relative tothe first analog-to-digital converter. The method further includes:suppressing noise via a decimation filter of the secondanalog-to-digital converter; performing a delta-sigma conversion via thesecond analog-to-digital converter to generate a second digital signalbased on the analog output of the first analog-to-digital converter,where the analog output of the first digital-to-analog converter isgenerated based on the analog input signal; and combining the firstdigital signal and the second digital signal to provide a resultantoutput signal.

In other features, an analog-to-digital converter is provided andincludes a digital-to-analog converter circuit, a sample and holdcircuit, a subtractor, an amplifier, a latch and a successiveapproximation module. The digital-to-analog converter circuit isconfigured to convert multiple bits of a digital signal to an analogsignal. The sample and hold circuit is configured to sample an analoginput signal. The subtractor is configured to subtract the analog signalfrom an output of the sample and hold circuit. The digital-to-analogconverter circuit includes: a first digital-to-analog converterconfigured to convert a first most-significant-bit of the bits; and asecond digital-to-analog converter configured to convert a firstleast-significant-bit of the bits, where the second digital-to-analogconverter is a delta-sigma digital-to-analog converter. The amplifier isconfigured to amplify an output of the digital-to-analog convertercircuit. The latch is configured to latch an output of the amplifier.The successive approximation module is configured to (i) receive anoutput of the latch, and (ii) perform successive approximations togenerate the digital signal.

In other features, an analog-to-digital converter is provided andincludes a sample and hold circuit, a first analog-to-digital converter,a second analog-to-digital converter and a combination circuit. Thesample and hold circuit is configured to sample an analog input signalto generate multiple bits. The first analog-to-digital converter isconfigured to generate a first digital signal based on the analog inputsignal. The first analog-to-digital converter includes a charge-sharingdigital-to-analog converter and a charge redistributiondigital-to-analog converter. The charge-sharing digital-to-analogconverter is configured to convert a first most-significant-bit of theplurality of bits. The charge redistribution digital-to-analog converteris configured to convert a first least significant bit of the bits. Thefirst digital signal is generated based on an output of thecharge-sharing digital-to-analog converter and an output of the chargeredistribution digital-to-analog converter. The second analog-to-digitalconverter is configured to generate a second digital signal based on anoutput of the first analog-to-digital converter. The secondanalog-to-digital converter includes a delta sigma digital-to-analogconverter. The delta sigma digital-to-analog converter is configured toconvert a second least significant bit of the bits. The second digitalsignal is generated based on an output of the delta sigmadigital-to-analog converter. The second analog-to-digital converter is afine conversion analog-to-digital converter relative to the firstanalog-to-digital converter. The combination circuit is configured tocombine the first digital signal and the second digital signal toprovide a resultant output signal.

In other features, a method is provided and includes: sampling an analoginput signal to generate multiple bits; and generating a first digitalsignal based on the analog input signal via a first analog-to-digitalconverter. The generation of the first digital signal includesconverting a first most-significant-bit of the bits via a charge-sharingdigital-to-analog converter, and converting a first least significantbit of the bits via a charge redistribution digital-to-analog converter.The first digital signal is generated based on an output of thecharge-sharing digital-to-analog converter and an output of the chargeredistribution digital-to-analog converter. The method further includesgenerating a second digital signal based on an output of the firstanalog-to-digital converter via a second analog-to-digital converterincluding converting a second least significant bit of the bits via adelta sigma digital-to-analog converter. The second digital signal isgenerated based on an output of the delta sigma digital-to-analogconverter. The second analog-to-digital converter is a fine conversionanalog-to-digital converter relative to the first analog-to-digitalconverter. The method further includes combining the first digitalsignal and the second digital signal to provide a resultant outputsignal.

In other features, an analog-to-digital converter is provided andincludes a digital-to-analog converter circuit, a sample and holdcircuit, a subtractor, an amplifier, a latch and a successiveapproximation module. The digital-to-analog converter circuit isconfigured to convert bits of a digital signal to an analog signal. Thesample and hold circuit is configured to sample an analog input signal.The subtractor is configured to subtract the analog signal from anoutput of the sample and hold circuit. The digital-to-analog convertercircuit includes: a first digital-to-analog converter configured toconvert a first most-significant-bit of the bits; a seconddigital-to-analog converter configured to convert a first leastsignificant bit of the bits; and a third digital-to-analog converterconfigured to convert a second least-significant-bit of the bits. Theamplifier is configured to at least one of amplify or integrate anoutput of the digital-to-analog converter circuit. The latch isconfigured to latch an output of the amplifier. The successiveapproximation module is configured to (i) receive an output of thelatch, and (ii) perform successive approximations to generate thedigital signal.

Further areas of applicability of the present disclosure will becomeapparent from the detailed description, the claims and the drawings. Thedetailed description and specific examples are intended for purposes ofillustration only and are not intended to limit the scope of thedisclosure.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block schematic view of an example SAR-ADC incorporating ahybrid SAR-DAC in accordance with an embodiment of the presentdisclosure.

FIG. 2 is a schematic view of an example of the hybrid SAR-DAC of FIG.1.

FIG. 3 illustrates an example of an analog-to-digital conversion methodincluding a digital-to-analog conversion method which in accordance withan embodiment of the present disclosure is implemented by the SAR-ADC ofFIG. 1.

FIG. 4 is a block schematic view of an example of a SAR-ΔΣ ADCincorporating coarse and fine DACs in accordance with an embodiment ofthe present disclosure.

FIG. 5 is a block schematic view of examples of a SAR comparator and aΔΣ comparator of the SAR-ΔΣ ADC of FIG. 4.

FIG. 6 is an example plot of SAR residual voltage ranges in accordancewith an embodiment of the present disclosure.

FIG. 7 is another example plot of SAR residual voltage ranges inaccordance with an embodiment of the present disclosure.

FIG. 8 is an example signal plot for the SAR-ΔΣ ADC of FIG. 4.

FIG. 9 is a block schematic view of an example of a SAR-ΔΣ ADCincorporating a switched integrator and latch for both coarse and fineDAC conversions in accordance with an embodiment of the presentdisclosure.

FIG. 10 illustrates another analog-to-digital conversion method inaccordance with an embodiment of the present disclosure is implementedby the SAR-ΔΣ ADC of FIG. 9.

FIG. 11 is a block schematic view of another example of a hybrid DACincluding a charge-sharing (CS) charge-redistribution (CR) segmented DACand a ΔΣ DAC in accordance with an embodiment of the present disclosure.

FIG. 12 illustrates an example of another analog-to-digital conversionmethod in accordance with an embodiment of the present disclosure isimplemented by the hybrid DAC of FIG. 11.

FIG. 13 is a block schematic view of an example of a 4-way interleavedSAR-ΔΣ ADC in accordance with an embodiment of the present disclosure.

FIG. 14 is an example signal plot and a timing diagram for the ADC ofFIG. 13.

In the drawings, reference numbers are reused to identify similar and/oridentical elements.

DESCRIPTION

The below disclosed examples provide different implementations ofSAR-DACs. These different implementations are referred to as “hybridcharge-sharing charge-redistribution SAR-DACs” or simply “hybridSAR-DACs” and have corresponding disclosed circuits, systems andmethods. The hybrid SAR-DACs are introduced to provide area efficientSAR-ADCs due to SAR-DAC architectures that minimize substrate surfacearea requirements of both a DAC core and a DAC reference voltagegenerator. Examples of hybrid SAR-DACs are shown in FIGS. 4, 6, 9, 11and 12.

FIG. 1 shows a SAR-ADC 130 that includes a hybrid SAR-DAC 132, anamplifier 134, a latch 136 and a SAR control module 138. A combinationof the amplifier 134 and the latch 136 referred to as a comparator and,in an embodiment, compares an output of the hybrid SAR-DAC 132 with, forexample, a reference voltage Vref. The amplifiers disclosed herein arereferred to as pre-amplifiers as the amplifiers perform amplificationprior to latching and successive approximation. The SAR control module138 is part of a feedback loop that feeds back a digital signalV_(DACIN). The hybrid SAR-DAC 132 receives an analog input voltageV_(ADCIN), the digital signal V_(DACIN), and a S/H phase signal φ_(SH).The hybrid SAR-DAC 132 includes a sample and hold (S/H) circuit 140, acharge-sharing (CS) DAC (CS-DAC) 142, and a charge-redistribution (CR)DAC (CR-DAC) 144. The S/H circuit 140 receives and samples the analoginput voltage V_(ADCIN). The DACs 142, 144 perform digital-to-analogconversions, as further described below. Output of the S/H circuit 140is provided to the CS-DAC 142. Output of the CS-DAC 142 is provided tothe CR-DAC 144. Output of the CR-DAC 144 is provided to the amplifier134.

The amplifier 134 provides an amplified output across a capacitor Ccmp.The latch 136 operates based on a clock signal Clk and latches an outputof the amplifier 134, which is provided to the SAR control module 138.The SAR control module 138 performs a successive approximation algorithm(or sometimes referred to as “a binary search algorithm”) based on theoutput of the latch 136 to provide a binary code, which is dependent on(i) a current bit being approximated and (ii) bits previouslyapproximated. The SAR control module 138 provides (i) the S/H phasesignal φ_(SH) to the S/H circuit 140, and (ii) the digital signalV_(DACIN) in the form of switch control signals b[1:n] to the DACs 142,144 to control switches of bit circuits of the DACs 142, 144.

FIG. 2 shows a hybrid SAR-DAC 150, which is a n-bit fully differentialSAR-DAC and in one embodiment replaces the hybrid SAR-DAC 132 of FIG. 1.The SAR-DAC 150 includes DACs 152, 154, which are binary-weighted. Abinary weighted DAC refers to a DAC where capacitance weighting for eachbit being converted is a product of (i) 2^(bit#-1) and (ii) acapacitance weighting for a corresponding bit (e.g., a most significantbit for a charge sharing DAC or a least significant bit (LSB) for acharge redistribution DAC). For example, if a capacitance weighting fora LSB is 600 atto-Farad (aF), then a capacitance weighting for a secondbit is 600·2²⁻¹=1200 aF, a third bit is 600·2³⁻¹=2400 aF, etc. . . . .Each of the DACs 152, 154 is binary-weighted as the value of thecapacitor pair C_(CSm1)=C_(CSm2) scales between the different bits (n .. . to p+1 for DAC 152 and p . . . to 1 for DAC 154) according to thelaw C_(CSm1)(j)=C_(CSm2)(j)=C0*2^(j) (orC_(CSm1)(j)=C_(CSm2)(j)=C0·2^(j)), where j is the bit number rangingfrom 1 to n. The capacitors C_(CSm1), C_(CSm2) are shared based oncontrol signals b_(n), b _(n). The capacitors C_(CRp1), C_(CRp2) areconnected to reference voltages based on control signals b_(n), b _(n).The hybrid SAR-DAC 150 includes (i) a m-bit binary weightedcharge-sharing (CS) DAC (CS-DAC) 152 that resolves a first mmost-significant-bits (MSBs), and (ii) a p-bit binary weightedcharge-redistribution (CR) DAC (CR-DAC) 154 that resolves last p LSBs,where n=m+p and m and p are integers. In one embodiment, the operationof the SAR-DAC 150 is applied to non-binary weighted DACs.

The CS-DAC 152 includes bit circuits p+1 to n, where n is the number ofbits being converted. Each of the bit circuits p+1 to n includes a firstcapacitor C_(CSm1), a first pair of switches 156, a second pair ofswitches 158, and a second capacitor C_(CSm2). In one embodiment, thecapacitors C_(CSm1), C_(CSm2) have the same capacitance and areconnected to a ground reference 160. A first node 162 is connected tothe first capacitor C_(CSm1) and receives a reference voltage V_(REFP)based on a state of a first switch 164. A second node 166 is connectedbetween one of the switches 156 and one of the switches 158. The secondnode 166 receives an input voltage VINP based on a state of a firstinput switch 168, and provides an output voltage VRP. A third node 170is connected between a second one of the switches 156 and a second oneof the switches 158. The third node 170 receives an input voltage VINNbased on a state of a second input switch 172 and provides an outputvoltage VRN. A fourth node 173 is connected to capacitor C_(CSm2), andreceives a reference voltage V_(REFN) based on a state of a switch 174.In one embodiment, a difference between the input voltages VINP, VINN isthe same as the input voltage V_(ADCIN) of FIG. 1. Each of the switches156, 158 receives a corresponding one of control signals b_(n), b _(n).Each of the switches 164, 168, 172, 174 is controlled by a S/H phasesignal φ_(SH), which part of the sample and hold circuit 140 of FIG. 1.The switches 164, 174 receive reference voltages V_(REFP), V_(REFN). Twosampling capacitors CS are connected in series between nodes 166, 170. Acommon mode voltage VCM exists between the capacitors CS. In oneembodiment, the common mode voltage VCM is predetermined and/orgenerated by a voltage generator and/or a control module (e.g., one ofthe control modules disclosed herein). Each of the bit circuits p+1 to nreceives the input voltages VINP, VINN, performs a respective conversionbased on corresponding ones of received control signals b_(n), b _(n)from the SAR control module 138 of FIG. 1 and provides output voltagesVRP, VRN.

A S/H circuit 175 includes the switches 168, 172 that receive the analoginput voltages VINP, VINN. The analog input voltages VINP, VINN areprovided to capacitors CS, which are connected to inputs of CS-DAC 152.A first pair of switches 176 is connected in series and between (i) afirst terminal connected between the switch 168 and a first one of thecapacitors CS, and (ii) a second terminal connected between the switch172 and a second one of the capacitors CS. A second pair of switches 178is connected in series and between (i) a first terminal connectedbetween the first one of the capacitors CS and a first input of theCS-DAC 152, and (ii) a second terminal connected between the second oneof the capacitors CS and a second input of the CS-DAC 152. Terminalsbetween the first pair of switches 176 and between the second pair ofswitches 178 are connected to ground. The switches 168, 172, 178 receiveS/H phase signal φ_(SH). The switches 176 receive an inverted version ofthe S/H phase signal φ_(SH).

The CR-DAC 154 is connected to the output of the CS-DAC 152 and includesbit circuits 1 to p. Each of the bit circuits 1 to p includes a firstpair of switches 180, a first capacitor C_(CRp1), a second capacitanceC_(CRp2), and a second pair of switches 182. The capacitors C_(CRp1),C_(CRp2) are the same capacitance, in an embodiment. The first pair ofswitches 180 are connected respectively between the voltage referencesV_(REFN), V_(REFP) and the first capacitor C_(CRp1). The second pair ofswitches 182 are connected respectively between the voltage referencesV_(REFN), V_(REFP) and the second capacitor C_(CRp2). Each of theswitches 180, 182 in the bit circuits 1 to p receives a correspondingone of digital output bits (or control signals) b_(n), b _(n) from theSAR control module 138 for the bit number associated with the bitcircuit of the switches 180, 182.

A first node 184 between the switches 180 and the capacitor C_(CRp1)receives the common mode voltage VCM based on a state of a first S/Hswitch 186. A second node 185 between the capacitor C_(CRp1) and theCS-DAC 152 provides the positive output voltage VRP. A third node 187between the capacitor C_(CRp2) and a second output of the CS-DAC 152provides the negative output voltage VRN. A fourth node 188 between theswitches 182 and the second capacitor C_(CRp2) receives the common modevoltage VCM based on a state of switch 190. The switches 186, 190receive the S/H phase signal φ_(SH). Each of the bit circuits 1 to p:receives the input voltages VINP, VINN; performs a respective conversionbased on corresponding ones of the control signals b_(n), b _(n); andprovides outputs voltages VRP, VRN.

During a sampling phase, capacitor CS and CCS_(DAC) of the CS-DAC 152are pre-charged with input voltages VINP, VINN and reference voltagesV_(REFP), V_(REFN) respectively while capacitor CCR_(DAC) of the CR-DAC154 is pre-charged at VCM. During the sampling phase, each of theswitches 164, 168, 172, 174, 186, 190 are closed and switches 156, 158,180, 182 are open. During a first m-cycles, states of the switches 156,158, 180, 182 change based on the control signals b_(n), b _(n) and thecharges are shared (additively/subtractively) in successiveapproximations between the capacitors CS and CCS_(DAC) to be minimizedat the end of the m-cycle conversion. Each of the m-cycles is associatedwith a respective successive approximation. A residual output voltageafter the m-cycles is then converted in successive approximation usingthe CR-DAC 154. An n-bit data output of the hybrid SAR-DAC 150 providedafter a last cycle n as represented by equation 1, where

$\frac{CS}{{CS} + {CCS}_{DAC} + {CCR}_{DAC}}$is the transfer function from the reference voltage to the input of thecomparator.

$\begin{matrix}{{{Vr}(n)} = {\frac{CS}{{CS} + {CCS}_{DAC} + {CCR}_{DAC}}{\quad{\left\lbrack {{- V_{IN}} + {\left( {\sum\limits_{i = 1}^{m}{b_{i + p}\frac{{CCS}_{i}}{CS}}} \right) \cdot V_{REF}} + {\left( {\sum\limits_{i = 1}^{p}{b_{i}\frac{{CCR}_{i}}{CS}}} \right) \cdot V_{REF}}} \right\rbrack\;,{{{w{here}}\mspace{14mu}{\sum\limits_{i = 1}^{m}{CCS}_{i}}} = {CCS}_{DAC}},{{\sum\limits_{i = 1}^{p}{CCR}_{i}} = {CCR}_{DAC}},{b_{i} = {\pm 1}}}}}} & (1)\end{matrix}$

The hybrid SAR-DAC 150 uses charge-sharing capacitors for MSBtransitions. Switch parasitics associated with the MSBs is a smallportion of the capacitance of the hybrid SAR-DAC 150 and does notsignificantly impact linearity performance of the hybrid SAR-DAC 150.Reference capacitors, capacitors connected to reference voltageterminals, do not have constraints on noise/ripple during the MSBstransitions, where the noise/ripple on the reference capacitors is at amaximum. Consequently, selection restrictions of the referencecapacitors is relaxed (or reduced) to reduce a required substratesurface area. Smaller reference capacitors are used due to the CS-DAC152 converting MSBs. Since smaller capacitances are used, surface areaneeded for the reference capacitors is reduced. Reference capacitors(not shown in FIG. 2) are used to minimize the ripple on the referencevoltages VREFP and VREFN due to the DAC activity (connecting anddisconnecting the DAC capacitances C_(CSm1), C_(CSm2), C_(CRp1),C_(CRp2), etc. . . . in FIG. 2).

The hybrid SAR-DAC 150 uses charge-redistribution capacitors for LSBtransitions. Noise/ripple of reference voltages during LSB transitionsis small and has a transfer function corresponding to comparator inputnodes with a large attenuation benefit due to the charge-sharingcapacitors associated with the MSBs, as indicated in equation 1.Consequently, the noise/ripple during the LSB transitions does notsignificantly impact linearity/noise performance of the hybrid SAR-DAC150. Capacitors associated with the LSBs are parasitic-insensitive(switch parasitic is on the reference sides of DAC capacitors) and theLSB capacitance is therefore scaled down to a technology node limitcapacitance (e.g., C_(min) _(_) _(tech)) or to a kT/C limit (e.g.,C_(min) _(_) _(noise)), which allows a DAC core area to be minimized.Alignment between the CS-DAC 152 and the CR-DAC 154 is accomplishedthrough calibration of the capacitors C_(CSm1), C_(CSm2), C_(CRp1),C_(CRp2). This calibration is performed to adjust and/or measure thevalues of the capacitors C_(CSm1), C_(CSm2), C_(CRp1), C_(CRp2) due todifferences between preselected capacitances and actual capacitances.

Referring again to FIG. 1, the SAR control module 138 includes a SAR andcontrol logic devices for generating, for example bit control signalsb_(n), b _(n), S/H phase signal φ_(SH), etc. The SAR control module 138performs a successive approximation algorithm to generate the bitcontrol signals b_(n), b _(n) based on a clock signal Clk and an outputof the latch 136. For further defined structure of the SAR controlmodule 138 of FIG. 1 see below provided methods and below provideddefinition for the term “module”. In one embodiment, the circuitsdisclosed herein are operated using example methods illustrated in FIGS.2, 7 and 14.

FIG. 3 shows an example of an analog-to-digital conversion methodincluding a digital-to-analog conversion method. Although the followingoperations are primarily described with respect to the implementationsof FIGS. 1-2, the operations are readily modified to apply to otherimplementations of the present disclosure. The operations areiteratively performed. In an embodiment, and begin at 200. At 202, theS/H circuit 140 receives the analog signal VINP, VINN. The followingoperations 204, 206 and 208 are performed during the same (first) periodof time. At 204, the S/H circuit 140 including switches 168, 172 samplesand holds a voltage of the analog signal VINP, VINN. At 206, capacitorsC_(CSm1), C_(CSm2) of the CS-DAC 152 are charged based on referencevoltages V_(REFP), V_(REFN). At 208, capacitors C_(CRp1), C_(CRp2) ofthe CR-DAC 154 are charged based on the common mode voltage VCM.

At 210, the hybrid DAC 150 performs a conversion for current bit duringa second period of time. If a MSB of a predetermined number of MSBs isbeing converted, then the SAR control module 138 at 210A generates thebit control signals b[1:n] such that charges on the capacitors C_(CSm1),C_(CSm2) of CS-DAC 152 are shared by changing states of switches 156,158. If a LSB of a predetermined number of LSBs is being converted, thenthe SAR control module 138 at 210B generates the bit control signalsb[1:n] such that charges on the capacitors C_(CRp1), C_(CRp2) areredistributed by changing states of the switches 180, 182 to connect thecapacitors C_(CRp1), C_(CRp2) to received reference voltages V_(REFN),V_(REFP).

At 212, output of the hybrid SAR-DAC 132 is provided to the amplifier134. The output is residual voltage Vr (or positive and negative outputresidual voltages VRP, VRN of the nodes 166, 170). At 214, the latch 136latches the residual voltage Vr based on the clock signal Clk. At 216,the SAR control module 138 performs a successive approximation algorithmbased on the clock signal Clk and latched amplified output of the latch136. The successive approximation algorithm includes, for example,converting an analog signal into a discrete digital representation via abinary search of all possible quantization levels before finallyconverging upon a digital output for each conversion.

At 218, the SAR control module 138 determines whether another cycle forcurrent conversion is to be performed. If another conversion is to beperformed, task 210 is performed, otherwise task 219 is performed. At219, the SAR control module 138 outputs a digital signal representing aword converted by the SAR-ADC 130.

At 220, the SAR control module 138 determines whether another conversionis to be performed. If another conversion is to be performed, task 222is performed, otherwise the method ends at 226. At 222, a conversioncount is incremented by the SAR control module 138. The method ends at226.

The above-described operations are meant to be illustrative examples; inone or more embodiments, the operations are performed sequentially,synchronously, simultaneously, continuously, during overlapping timeperiods or in a different order depending upon the application. Also, inone or more embodiments, one or more of the operations is not performedor skipped depending on the implementation and/or sequence of events.

In addition, to providing area efficient SAR-ADCs with reduced size ascompared to traditional SAR-ADCs, power efficient ADCs that consume lesspower than traditional ADCs are also disclosed herein. The disclosedADCs include successive approximation delta sigma (ΔΣ) ADCs. The ΔΣ ADCsare applicable to the hybrid SAR-DACs or used in other applicationsseparate and/or independent of the hybrid SAR-DACs. The ΔΣ ADCs areapplicable to any type of ADC, where thermal noise is a significantcontributor to an effective number of bits (ENOBs) being converted.

In one embodiment, a combined low power and substrate surface areaefficient ADC is used on a mixed signal system-on-chip (SOC). In scaledtechnologies, power challenges have been addressed using SARarchitectures often in combination with techniques such as redundancyand asynchronous operation and time interleaving to meet applicationsampling rate requirements. However, for high-resolution ADCs (e.g.,greater than or equal to 10 effective number of bits (ENOB)), atraditional SAR-ADC is intrinsically energy inefficient since theSAR-ADC reuses a same low noise comparator to perform both (i) coarseconversions where little accuracy is required, and (ii) fine conversionswhere thermal noise is of importance. The energy inefficiency associatedwith a traditional SAR-ADC is addressed by the example hybrid SAR-ΔΣADCs disclosed below with respect to FIGS. 4-6.

Substrate surface area associated with a SAR-ADC has been reduced inrecent years with the introduction of digital DAC linearity calibrationschemes that relax matching requirements allowing SAR-DAC capacitors tobe scaled down to kT/C limits. In high-resolution noise-limited DAC, DACcapacitance grows 4 x for every extra bit of resolution while, at thesame time, a size of voltage noise/ripple on a reference voltage needsto be reduced 2× to preserve linearity. This sets difficult to meetrequirements for a reference generator of a high-resolutioncharge-redistribution SAR-ADC, especially if no external components areused. This requires large on chip capacitors. Traditionally this issuehas been addressed using (i) traditional DAC switching schemes thatoptimize current absorbed, or (ii) traditional DAC topologies that aremore immune to reference voltage ripple (e.g., current steering orcapacitive charge sharing).

FIG. 4 shows an example ADC 250. The ADC 250 includes a SAR coarse ADC252, a ΔΣ fine ADC 254 and a recombination module 256. The SAR coarseADC 252 includes a S/H circuit 258, a subtractor 260, a SAR comparator262, a SAR control module 264, and a SAR-DAC 266. The ΔΣ fine ADC 254includes a subtractor 270, a ΔΣ comparator 272, a ΔΣ control module 273,a ΔΣ decimation filter 274, and a ΔΣ DAC 276. The S/H circuit 258receives an analog input voltage LANE_(IN), samples the analog inputvoltage LANE_(IN), and holds the sampled voltage based on a S/H phasesignal φ_(SH). The term “lane” as used herein refers to a signal path,bit circuit, and/or a conductor. In one embodiment, a lane has a singleinput and a single output or a differential input and a differentialoutput. The subtractor 260 subtracts output of the SAR-DAC 266 from anoutput of the S/H circuit 258.

The SAR comparator 262 compares an output of the subtractor 260 with areference voltage Vref based on an enable signal enSAR. The SAR controlmodule 264 performs a successive approximation algorithm and provides adigital output to the SAR-DAC 266. After a last cycle of ananalog-to-digital conversion, the SAR control module 264 provides adigital output SAR_(OUT) to the recombination module 256. In oneembodiment, the S/H circuit 258, subtractor 260 and SAR-DAC 266 arereplaced by one of the hybrid SAR-DACs 132, 150 of FIGS. 1 and 2. TheSAR-DAC 266 or one of the hybrid SAR-DACs 132, 150 performs adigital-to-analog conversion for a predetermined number of bits (e.g.,10 bits with 9 bit resolution). In one embodiment, if one of the hybridSAR-DACs 132, 150 is used, 4 MSBs and 6 LSBs are converted, where one ofthe CS-DACs 142, 152 converts the 4 MSBs and one of the CR-DACs 144, 154converts the 6 LSBs.

The subtractor 270 subtracts a residual output voltage Vr of the SARsubtractor 260 from an output of the ΔΣ-DAC 276. The ΔΣ comparator 272compares an output of the subtractor 270 with the reference voltage Vrefbased on an enable signal enΔΣ. The ΔΣ control module 273 performs a ΔΣalgorithm and provides a digital output to the ΔΣ-DAC 276 and the ΔΣdecimation filter 274. In one embodiment, the ΔΣ algorithm includes (i)providing an output of the comparator 272 directly to the ΔΣ DAC 276 anddirectly to the ΔΣ decimation filter 274, and (ii) tracking a number ofcycles performed by the ΔΣ fine ADC 254 to provide a status of aconversion being performed. The ΔΣ decimation filter 274 filters anoutput of the ΔΣ control module 273 and provides a digital output to therecombination module 256. After a last cycle of an analog-to-digitalconversion, the ΔΣ decimation filter 274 provides the digital outputΔΣZ_(OUT) to the recombination module 256. In one embodiment, the ΔΣ-DAC276 is configured as shown in FIG. 11 and performs a digital-to-analogconversion for one or more LSBs. As an example, the one or more LSBsinclude a first LSB. The next 6 LSBs are converted by the SAR-DAC 266 orone of the hybrid SAR-DACs 132, 150. The recombination module 256combines the outputs SAR_(OUT) and ΔΣ_(OUT) to provide a digital outputLANE_(OUT) (12-bit output with a 11-bit resolution).

During operation, the ΔΣ fine ADC 254 converts the SAR residual error(V_(S)−V_(DAC)) at an end of the conversion performed by the SAR coarseADC 252. The SAR coarse ADC 252 periodically and/or iteratively performsmultiple conversion cycles for each conversion performed by the ΔΣ fineADC 254. In one embodiment, the SAR control module 264 performs multiplesuccessive approximations until settling on a final successiveapproximation for one or more bits prior to the ΔΣ fine ADC 254performing a conversion based on a difference between the result of thefinal successive approximation and an output of the ΔΣ DAC 276. Thefiltered output ΔΣ_(OUT) of the ΔΣ fine ADC 254 is combined with theoutput SAR_(OUT) of the SAR conversion performed by the SAR coarse ADC252 to provide a final output LANE_(OUT). During a SAR phase, thereceived signal LANE_(IN) is sampled and the SAR coarse ADC 252 performsa first N conversions, where enSAR=1 and enSD=0 for N clock periods.Subsequent to the SAR phase, a ΔΣ phase is performed, the error(V_(S)−V_(DAC)) at the end of the SAR phase is fed into the ΔΣ fine ADC254 and is converted for M clock periods, where enSAR=0 and enSD=1 forthe M clock periods.

FIG. 5 shows examples of the SAR comparator 262 and ΔΣ comparator 272.The SAR comparator 262 includes an amplifier 280, a first latch 282 anda first AND gate 284. The amplifier 280 (i) receives the residualvoltage Vr from the subtractor 260 of FIG. 4, and (ii) compares theresidual voltage Vr to a reference voltage or ground reference, asshown. The amplifier 280 receives the voltage Vr and a reference voltagefrom, for example, a reference terminal or ground, as shown. Theamplifier 280 compares the voltage Vr to the reference voltage. Anoutput of the amplifier 280 is connected to a capacitor Ccmp, which isconnected to a ground reference. The latch 282 latches (or holds) avoltage seen across the capacitor Ccmp based on an output of the firstAND gate 284 and provides the voltage as an output. The first AND gate284 receives the enable signal enSAR and clock signal Clk. The SARcontrol module 264 of FIG. 4 generates the enable signal enSAR, asdescribed above with respect to FIG. 4.

The ΔΣ comparator 272 includes a loop filter 286, a second latch 288 anda second AND gate 290. The loop filter 286 includes and/or beimplemented as an integrator and receives an output of the subtractor270 (shown in FIG. 4). The loop filter 286 filters and/or integrates theoutput of the subtractor 270. The integration includes summing outputsof the subtractor 270. An output of the loop filter 286 is provided tothe second latch 288, which is controlled by the second AND gate 290.The second AND gate 290 receives the enable signal enΔΣ and the clocksignal Clk.

Referring again to FIG. 4, the SAR coarse ADC 252 is a non-binary 10-bitSAR-ADC (with 9 bit equivalent resolution). The ΔΣ fine ADC 254 isimplemented as a single-bit first-order continuous-time incremental ΔΣADC (with a 2× over range). The SAR coarse ADC 252 and A fine ADC 254 asshown provide improved resolution over traditional SAR-ADCs due to useof both a SAR coarse ADC and a ΔΣ fine ADC having a 2× over range. Thisis shown in FIG. 6, where the ΔΣ fine ADC input range (or input range ofthe ΔΣ fine ADC 254) is twice the SAR quantization range (or outputrange of the SAR coarse ADC 252). FIG. 6 shows an example plot of SARresidual voltage ranges for a ΔΣ fine ADC input, SAR quantization plusSAR loop noise, and SAR quantization alone. The SAR residual voltageV_(SAR Residual) (or Vr) is equal to V_(S)−V_(DAC).

FIG. 7 shows an example plot of SAR voltage residual ranges for anotherexample implementation, where the ΔΣ fine ADC input range (or full scalevoltage V_(FSSD) of the ΔΣ fine ADC 254) is greater than 3 times SARloop noise σ_(loop). By setting the full scale voltage V_(FSSD) greaterthan 3 times SAR loop noise φ_(loop), noise and power requirementspermitted to be reduced.

FIG. 8 shows an example signal plot illustrating cycle timing of theSAR-ΔΣ ADCs. A clock signal Clk, a S/H phase signal φ_(SH), a SAR (orfirst) enable signal enSAR, and a ΔΣ (or second) enable signal areshown. As shown, an analog input signal is sampled during an OFF periodT_(TH) of the clock signal Clk. Then the clock signal Clk is ON and thefirst enable signal enSAR is transitioned HIGH to enable the SARcomparator 262 and perform MSB conversions during a second periodT_(SAR). As an example, this occurs for 10 clock cycles. Following thesecond period, the first enable signal enSAR is transitioned LOW and thesecond enable signal enΔΣ is transitioned HIGH to enable the ΔΣcomparator 272. The ΔΣ comparator 272 is HIGH for a third period T_(ΔΣ)(e.g., 8 clock cycles) to perform LSB conversions. The overall time toperform an analog-to-digital conversion, which includes a sum of theperiods T_(TH), T_(SAR), T_(ΔΣ) is referred to as lane period T_(LANE).

After sampling of analog signal and a coarse SAR-ADC conversion phase(e.g., 10 clock cycles) of the SAR coarse ADC 252, a SAR residual erroris converted by the fine ΔΣ fine ADC 254 during a fine ADC conversionphase (e.g., 8 clock cycles). The decisions (e.g., 8 decisions)performed by the comparator of the ΔΣ fine ADC 254 are summed withdifferent weights before being recombined with the SAR_(ouT) to providea final code (e.g., the 12-bit signal LANE_(OUT)). In one embodiment,the operation of the ΔΣ fine ADC 254 is equivalent to applying an 8-taplow pass finite impulse response (FIR) filter and then performing adecimation by 8.

Once SAR_(OUT) and ΔΣ_(OUT) are combined, the residual error includes,in addition to sampling kT/C noise, only ΔΣ fine ADC 204 noisecomponents (i.e. ΔΣ quantization noise, latch thermal-noise, and loopfilter thermal-noise). All these components are low-pass filtered by theΔΣ decimation filter 274 (shown in FIG. 4), which in one embodimentoperates as a low pass FIR filter having an equivalent bandwidthapproximately equal to 1/TΔΣ, where TΔΣ is a conversion time of the ΔΣfine ADC 254. Since both quantization and latch thermal-noise are 1^(st)order shaped, the quantization and latch thermal-noise are stronglysuppressed by the FIR leaving as dominant terms the sampling kT/C noiseand the input referred noise of the integrator of the ΔΣ fine ADC 254with an integration time TAX. Since the integrator is inside a ΔΣ loop,a gain calibration is not needed and improved linearity performance isprovided.

Reduced Hardware Implementation

In one embodiment, the ΔΣ fine ADC 254 is embedded in and/or merged withthe SAR coarse ADC 252 and a single comparator latch is reused tominimize hardware overhead. FIG. 9 shows a SAR-ADC 300 that includes aS/H circuit 302, a subtractor 304, a comparator static amplifier (orgain integrator (Gmint)) 306, a latch 308, a SAR-ΔΣ control module 312,and a SAR-ΔΣ DAC 314. The SAR-ADC 300 is a reduced hardware example ofthe SAR-ΔΣ ADC of FIG. 4 with reduced number of hardware components andthus reduced overall size. If a switched integrator (e.g., switchedintegrator 315) is used as a SAR amplifier (e.g., replaces the amplifier280 of FIG. 5), then the switched integrator is reused as a first orderloop filter of a ΔΣ-ADC, as provided by a combination of the gainintegrator 306 and the switch 316. In one embodiment, the switchintegrator 315 includes and/or functions as a combination of the gainintegrator, a load capacitor Cint and/or the switch 316. When theswitched integrator is reused, the load capacitor Ccmp is not reset viaswitch 316. In one embodiment, the SAR comparator (e.g., combination ofamplifier 280 and latch 282 or combination of gain integrator 306 andlatch 308) is reused for, as an example, a single bit implementation,where a single bit is converted during each clock cycle.

The S/H circuit 302 receives an analog input voltage LANE_(IN), samplesthe analog input voltage LANE_(IN), and holds the sampled voltage basedon a S/H phase signal φ_(SH). The subtractor 304 subtracts an output ofthe SAR-ΔΣ DAC 314 from an output of the S/H circuit 302. The gainintegrator 306 receives an output voltage V_(SARRES) from the subtractor304 performs integration to provide an output across the load capacitorCint. A transistor 316 is connected across the load capacitor Cint andreceives a control input from an output of a NOR-gate 318. The NOR-gate318 receives an enable signal enΔΣ and a clock signal Clk, such that theswitch is On when both the enable signal enΔΣ and the clock signal Clkare low. The latch 308 latches, based on the clock signal Clk, a voltageacross the load capacitor Cint and the transistor 316. The SAR-ΔΣcontrol module 312 performs a ΔΣ algorithm and provides signal SAR_(OUT)to the recombination module 256. Digital output of the SAR-ΔΣ controlmodule 312 is provided to the SAR-ΔΣ DAC 314 and a ΔΣ decimation filter320. At the end of a conversion, the ΔΣ decimation filter 320 provides adigital output signal to the recombination module 256, which providesthe signal LANE_(OUT) (e.g., 12-bits representing analog input voltageprovided by signal LANE_(IN)). The latch 308, switch integrator 315, NORgate 318 operate as a combined coarse/fine ADC. The SAR-ΔΣ DAC 314replaces and operates similar as the SAR DAC 266 and ΔΣ DAC of FIG. 4.

Reconfiguring a comparator static amplifier Gmint into a ΔΣ integratorincludes stopping reset of the load capacitor Cint after a last SARcycle (an example of which is shown in FIG. 11). In one embodiment,timing signals used for the ΔΣ conversion are generated using aself-timed loop technique including, for example a delay lock loop togenerate timing signals, such as the S/H phase signal φ_(SH), the clocksignal Clk, and/or one or more enable signals (e.g., one of the enablesignals enΔΣ, enSAR shown in FIGS. 4 and 9).

A SAR-ADC that includes a CS-DAC (e.g., one of the CS-DACs 142, 152 ofFIGS. 1-2) reduces substrate surface area (and power) requirements of areference generator. A CS-DAC is highly insensitive to noise/ripple onvoltage references since the voltage references (e.g., voltagereferences V_(REFN), V_(REFP) of FIG. 2) are physically disconnected viaswitches (e.g., switches 164, 174 of FIG. 2) from the CS-DAC during asuccessive approximations phase, as described above. This allows for areduction in capacitances corresponding to voltage references. TheCS-DAC, however, requires a top plate switch (switches are on acomparator end of the CS-DAC 152, such as switches 156, 158 of FIG. 2)and thus is parasitic capacitance sensitive. Scaling of LSB capacitancesis therefore dictated by the parasitic capacitance of a minimum size topplate switch instead of kT/C noise requirements. For deep-submicrontechnology systems, this results in a higher DAC core area with respectto a CR-DAC. Use of a CS-DAC in combination with a CR-DAC allows for theDAC core area to not be increased since as described above the switchparasitics associated with MSBs is a small portion of the capacitance ofthe hybrid SAR-DAC. A large portion of the capacitance of the hybridSAR-DAC is associated with the CS-DAC. Capacitances associated with theLSBs, which are handled by the CR-DAC, are small. Thus, the overallcapacitance of the SAR-DAC is minimized, thereby reducing DAC core areaand/or corresponding substrate area.

Loop Noise Comparison

A conventional SAR ADC experiences loop noise generated by a switchedintegrator (efficient amplifier implementation) with a settling timeequal to T_(clk)/20, where the SAR loop noise is represented by equation2, g_(M,A) is the transconductance of the switched integrator, T_(settl)is settling time, T_(clk) is, k is Boltzman constant, T is samplingtime.

$\begin{matrix}{\sigma_{{loop},{SAR}} = {\sqrt{\frac{2{kT}}{g_{M,A}T_{settl}}} = \sqrt{\frac{40{kT}}{g_{M,A}T_{clk}}}}} & (2)\end{matrix}$

In one embodiment, the ADC architectures of FIGS. 4, 5 and/or 9 having afirst order single bit g_(m,lpf)-C type integrator circuit thatexperiences loop noise σ_(SD) represented by equation 3, where:g_(M,lpf) is gain associated with the integrator and low pass filter(provided by the capacitor C) of the integrator circuit;

$\frac{\pi^{2}V_{FSSD}^{2}}{9\; M^{3}}$is SD quantization noise, and

$\frac{2{KT}}{g_{M,{lpf}}{MT}_{clk}}$is loop filter thermal noise.

$\begin{matrix}{\sigma_{SD} = \sqrt{\frac{\pi^{2}V_{FSSD}^{2}}{9M^{3}} + \frac{2{KT}}{g_{M,{lpf}}{MT}_{clk}}}} & (3)\end{matrix}$By choosing VFSSD=3σ_(loop,SAR) and g_(M,lpf)=g_(M,A), the SAR loopnoise σ_(SD) is represented by equation 4.

$\begin{matrix}{\sigma_{SD} \cong {\sigma_{{loop},{SAR}}\sqrt{\frac{\pi^{2}}{M^{3}} + \frac{1}{20M}}}} & (4)\end{matrix}$

If M=4, the SAR amplifier noise is suppressed by approximately 8decibels (dB) with a 1.5 bit increase in ENOB with 4 additional clockcycles.

Energy Efficiency

A traditional SAR ADC, using a same amplifier for each SAR conversioncycle, having MSB decisions done with full noise performance, andimplementing an integrator with a limited settling time T_(settl) (e.g.,T_(clk)/20), in one embodiment, has a required amount of amplifier powerrepresented by equation 5.

$\begin{matrix}{P_{SAR} \propto {\frac{2{KT}}{\sigma_{loop}^{2}}N_{SAR}}} & (5)\end{matrix}$

The SAR amplifier noise for N_(b) SAR conversion cycles of the ADCarchitectures of FIGS. 4-5 and/or 9 is suppressed during the ΔΣ phase.The SAR amplifier (e.g., the amplifier 280 of FIG. 5) is noisier andrequires less power than the amplifier of the traditional SAR ADC. Loopfilter noise is filtered by the ΔΣ decimation filter, where bandwidth(BW)=1/MT_(clk). In one embodiment, the power provided to the SARamplifiers of the disclosed ADC architectures is represented by equation6.

$\begin{matrix}\left. {P_{{SAR}\text{-}{SD}} \propto {\frac{2{kT}}{\sigma_{loop}^{2}}\left( {\frac{\pi^{2}}{M^{3}} + \frac{1}{20\; M}} \right)\left( {N_{b} + {20\; M}} \right)}}\rightarrow\frac{2{kT}}{\sigma_{loop}^{2}} \right. & (6)\end{matrix}$

for high M

There is a 3 times power reduction for a same ENOB, where M=8, from thetraditional SAR ADC and the disclosed ADC (or SAR-ΔΣ arrangement). Forthe same amount of noise and an example embodiment, the energy drawn bythe disclosed SAR-ΔΣ arrangement is equal to the energy used by thetraditional SAR ADC for a single SAR conversion cycle.

Energy efficiency is improved by incorporating a coarse ADC and aDelta-Sigma (ΔΣ) ADC as a fine ADC in a SAR-ADC as above-described.Energy efficiency is also improved while substrate surface arearequirements are significantly reduced by using a segmentedcharge-sharing charge-redistribution DAC, such as that described abovewith respect to FIGS. 1-2.

Referring again to FIG. 5, noise associated with the latches 282, 288(referred to as latch noise Vn) exists and/or is effectively provided(i) between the amplifier 280 and the latch 282, and (ii) between theloop filter 286 and the latch 288. The latch noise Vn is added to asignal provided to the latch 282 divided by dynamic gain of theamplifier 280. The latch noise Vn is added after the loop filter 286 andis suppressed by the feedback loop provided by the ΔΣ-DAC 276 of FIG. 4.

FIG. 10 shows an analog-to-digital conversion method. Although thefollowing operations are primarily described with respect to theimplementations of FIGS. 4-5 and 9, the operations are easily modifiedto apply to other implementations of the present disclosure. Theoperations are iteratively performed in an embodiment.

The method begins at 330. At 332, an analog signal LANE_(IN) isreceived. At 334, a S/H circuit (e.g., one of the S/H circuits 258, 302)receives the analog signal LANE_(IN).

The following operations 336-341 are performed by the SAR coarse ADC 252or the ADC 300. At 336, the subtractor 260, 304 subtracts an output ofthe DAC 266, 314 from a held value provided by the S/H circuit. At 337,the output Vr of the subtractor 260, 304 is compared with a referencevoltage, integrated, amplified and/or latched, as described above.

At 338, the control module 264, 312 executes a successive approximationalgorithm based on the output of the latch 282, 308 to generate adigital approximation signal. At 339, the control module 264, 312determines whether another cycle is to be performed for a current bit.If another cycle is to be performed, task 340 is performed, otherwiseoperations 341 and 342 are performed. At 340, the DAC 266, 314 performsa digital-to-analog conversion to generate an analog signal based on thedigital approximation signal. Task 336 is performed subsequent to task340. At 341, the last generated digital approximation signal is outputfrom the control module 264, 312 to the recombination module 256.

The following operations 342-346 are performed by the ΔΣ fine ADC 254 orthe hybrid ADC 300. At 342, the subtractor 270 subtracts an output ofthe ΔΣ-DAC 276 from Vr or the subtractor 304 subtracts the output of theSAR-ΔΣ DAC 314 from the held value of the S/H circuit 302. At 343, theoutput of the subtractor 270, 304 is compared with a reference voltage,integrated, amplified and/or latched, as described above.

At 344, the ΔΣ control module 273, 312 executes a ΔΣ algorithm based onthe output of the latch 288, 308 to generate a second digitalapproximation signal. At 345, the ΔΣ decimation filter 274, 320 filtersthe output of the ΔΣ control module 273, 312 to generate a digitalapproximation signal. At 346 the control module 264, 312 determineswhether another cycle is to be performed for a current bit. If anothercycle is to be performed, task 347 is performed, otherwise operation 348is performed. At 347 the ΔΣ-DAC 276 or SAR-ΔΣ DAC 314 (hybrid DAC)performs a digital-to-analog conversion of the digital approximationsignal generated at 345. Task 343 is performed subsequent to task 347.

At 348 the last generated digital approximation signal generated by theΔΣ fine ADC 254 or the hybrid ADC 300 is output from the ΔΣ decimationfilter 274 or the control module 312 to the recombination module 256. At349, the recombination module 256 combines the first converted digitalsignal and the second converted digital signals received at 341 and 348to generate a resultant output signal LANE_(OUT). As an example, themethod ends at 350.

The above-described operations are meant to be illustrative examples; inone or more embodiments, the operations are performed sequentially,synchronously, simultaneously, continuously, during overlapping timeperiods or in a different order depending upon the application. Also, inone embodiment, one or more of the operations are not performed orskipped depending on the implementation and/or sequence of events.

To exploit the reduced surface area advantage of a CS-DAC in selectionof voltage references while maintaining and/or minimizing the DAC corearea, a segmented SAR-DAC architecture is disclosed. A detailed exampleof this architecture is shown in FIG. 11. FIG. 11 shows a SAR-ADC 351that includes a S/H circuit 352, a hybrid DAC 354, a comparator staticpre-amplifier 356, a latch 358, and a SAR-ΔΣ control module 360. Thehybrid DAC 354 includes a CS-DAC 362, a CR-DAC 364 and a ΔΣ-DAC 366.Each bit circuit of the ΔΣ-DAC 366 is configured similarly to each bitcircuit of the CR-DAC 364, but with different capacitance weighting, asis further described below.

The S/H circuit 352 includes switches 370, 372 that receive analog inputvoltages LANE INP, LANE INN. The analog input voltages LANE INP, LANEINN are provided to capacitors CTH, which are connected to inputs ofCS-DAC 362. A first pair of switches 374 is connected in series andbetween (i) a first terminal connected between the switch 370 and afirst one of the capacitors CTH, and (ii) a second terminal connectedbetween the switch 372 and a second one of the capacitors CTH. A secondpair of switches 376 is connected in series and between (i) a firstterminal connected between the first one of the capacitors CTH and afirst input of the CS-DAC 362, and (ii) a second terminal connectedbetween the second one of the capacitors CTH and a second input of theCS-DAC 362. Terminals between the first pair of switches 374 and betweenthe second pair of switches 376 are connected to ground. The switches370, 372, 376 receive S/H phase signal φ_(SH). The switches 374 receivean inverted version of the S/H phase signal φ_(SH).

The CS-DAC 362 includes bit circuits 379 (e.g., bit circuits for bitsb[n:p+1] or b[10:7] as shown in FIG. 2) and is configured similar to theCS-DAC 152 of FIG. 2. Each of the bit circuits 379 includes capacitorsCS₁₀, a first pair of switches 380, and a second pair of switches 382.In one embodiment, the capacitors CS₁₀ have the same capacitance and areconnected to a ground reference 384. A first node 386 is connected toone of the capacitors CS₁₀, and receives a reference voltage V_(REFP)based on a state of a switch 388. A second node 390 is connected betweenone of the switches 380 and one of the switches 382. The second node 390receives an input voltage from one of the capacitors CTH and provides anoutput voltage to the CR-DAC 364. A third node 392 is connected betweena second one of the switches 380 and a second one of the switches 382.The third node 392 receives an input voltage from a second one of thecapacitors CTH and provides an output voltage to the CR-DAC 364. Asecond one of the switches 380 and a first one of the switches 382receives control signals b_(SAR10). A first one of the switches 380 anda second one of the switches 382 receives inverted control signal b_(SAR10) represented by inverter signals on certain ones of the switches380, 382. A fourth node 394 is connected to one of the capacitors CS₁₀,and receives a reference voltage V_(REFN) based on a state of a switch396. The switches 388, 396 are respectively controlled by a S/H phasesignal φ_(SH). The switches 388, 396 receive reference voltagesV_(REFP), V_(REFN). Bit circuits 379 respectively receive voltagesoutput from the S/H circuit 352 and perform a respective conversionbased on corresponding ones of received control signals b_(SAR10), b_(SAR10) from the SAR-A control module 360; and provides output voltagesto the comparator static pre-amplifier 356.

The CR-DAC 364 is connected to the output of the CS-DAC 362 and includesbit circuits 400 and is configured similar to the CR-DAC 154 of FIG. 2.Each of the bit circuits 400 includes a first pair of switches 402,capacitors CR₆, and a second pair of switches 404. In one embodiment,the capacitors CR₆ have the same capacitance. The first pair of switches402 is connected respectively between the voltage references V_(REFN),V_(REFP) and one of the capacitors CR₆. The second pair of switches 404is connected respectively between the voltage references V_(REFN),V_(REFP) and a second of the capacitors CR₆. Each of the switches 402,404 receives a corresponding one of control signals b_(SAR6), b _(SAR6)from the SAR-AE control module 360. As an example, the bit circuits 400receive respective control signals for converting bits [6:1].

A first node 410 between the switches 402 and a first terminal of thefirst one of the capacitors CR₆ is connected to the ground reference 384based on a state of a switch 412. A second node 414 connected to asecond terminal of the first one of the capacitors CR₆ provides a firstoutput voltage to the ΔΣ-DAC 366. A third node 416 connected to a firstterminal of the second one of the capacitors CR₆ provides a secondoutput voltage to the ΔΣ-DAC 366. A fourth node 418 between the switches404 and a second terminal of the second one of the capacitors CR₆ isconnected to the ground reference 384 based on a state of switch 420.The switches 412, 420 receive the S/H phase signal φ_(SH). Each of thebit circuits 400: receives input voltages from the CS-DAC 362; performsa respective conversion based on corresponding ones of the controlsignals b_(SAR6), b _(SAR6); and provides output voltages to thecomparator static pre-amplifier 356.

The input referred thermal noise is reduced by the architecture of theΔΣ-DAC 366, as described above with the ΔΣ ADC 254 of FIG. 4. The ΔΣ-DAC366 includes one or more bit circuits (a single bit circuit 430 isshown). The bit circuit 430 is connected to the output of the CR-DAC 364and includes a first pair of switches 432, capacitors CR_(ΔΣ), and asecond pair of switches 434. In one embodiment, the capacitors CR_(ΔΣ)have the same capacitance. The first pair of switches 432 is connectedrespectively between the voltage references V_(REFN), V_(REFP) and oneof the capacitors CR_(AE). The second pair of switches 434 is connectedrespectively between the voltage references V_(REFN), V_(REFP) and asecond of the capacitors CR_(ΔΣ). Each of the switches 432, 434 receivesa corresponding one of control signals b_(ΔΣ), b _(ΔΣ) from the SAR-ΔΣcontrol module 360. As an example, the bit circuit 430 receives acontrol signal for converting bit [1:0].

A first node 440 between the switches 432 and a first terminal of thefirst one of the capacitors CR_(ΔΣ) is connected to the ground reference384 based on a state of a switch 442. A second node 444 connected to asecond terminal of the first one of the capacitors CR_(ΔΣ) provides afirst output voltage to the ΔΣ-DAC 366. A third node 446 connected to afirst terminal of the second one of the capacitors CR_(ΔΣ) provides asecond output voltage to the ΔΣ-DAC 366. A fourth node 448 between theswitches 434 and a second terminal of the second one of the capacitorsCR_(ΔΣ) is connected to the ground reference 384 based on a state ofswitch 450. The switches 442, 450 receive the S/H phase signal φ_(SH).The bit circuit 430: receives input voltages from the CR-DAC 364;performs a respective conversion based on corresponding ones of thecontrol signals b_(ΔΣ), b _(ΔΣ); and provides output voltages to thecomparator static pre-amplifier 356.

The comparator static pre-amplifier 356 receives output voltages fromthe DACs 362, 364, 366 and performs integration to provide an outputacross a load capacitor Cint, as described above with respect to theintegrator 306 of FIG. 9. In one embodiment, the integration includessumming a difference between the output voltages from the DACs 362, 364,366 for a predetermined period of time. A transistor 460 is connectedacross the load capacitor Cint and receives a control input from anoutput of a NOR-gate 462. The NOR-gate 462 receives an enable signalenΔΣ and a clock signal Clk. The latch 358 latches, based on the clocksignal Clk, a voltage across the load capacitor Cint and the transistor460. Although the SAR-ΔΣ control module 360 is shown as including a ΔΣdecimation filter 464, in another embodiment the ΔΣ decimation filter464 is separate from the SAR-ΔΣ control module 360, as similarly shownin FIG. 9. The SAR-ΔΣ control module 360 provides: S/H phase signalφ_(SH) to the S/H circuit 352 and the DACs 362, 364, 366; a digitaloutput and signals φ_(SH), b _(SAR10), b_(SAR6), b _(SAR6), b_(ΔΣ), b_(ΔΣ) to the DACs 362, 364, 366; and signals enΔΣ, Clk to the NOR gate462. At the end of a conversion, the SAR-ΔΣ control module 360 providesa digital output as LANE_(OUT) (e.g., 12-bits representing analog inputvoltage provided by input voltage signals LANE INP, LANE INN).

The segmented architecture of the hybrid DAC 354 includes the CS-DAC 362for a predetermined number of MSBs (e.g., 4-MSBs), the CR-DAC 364 forLSBs (e.g., 6-LSBs) and/or the ΔΣ-DAC 366 for one or more LSBs. In oneembodiment, the ΔΣ-DAC 366 is not included. Switch parasiticcapacitances in the CS-DAC 362 are a small portion of the DACcapacitances of the hybrid DAC 354 and do not significantly impactlinearity performance. In one embodiment, the capacitances of the CR-DAC364 are scaled down as in a conventional CR-DAC. The voltage referencenoise/ripple for the LSBs is reduced by 2⁴ (or 2 to the power 4 or thenumber of MSBs). The voltage reference noise/ripple is also attenuatedat an input of the corresponding comparator due to the inclusion of thecapacitors CS, which are connected to the ground reference 384. As aresult and as an example, for a SAR-DAC core capacitance of 300femtoFarad (fF), capacitance of the corresponding reference lane is only5 picoFarad (pF).

FIG. 11 further includes a table including TH and DAC weights, whichrefer to respective capacitances of the S/H circuit 352 and DACs 362,364, 366. Each of the weights multiplied by, a predetermined capacitance(e.g., 600 atto-farad (aF)) provides the capacitance of thecorresponding capacitance. Multiple weights are provided for the DACs362, 364 respectively and correspond to bits converted by the DACs 362,364. For example, bit 7 of the CS-DAC 362 has the weight 37, which ifmultiplied by the predetermined capacitance provides the value of eachof the capacitors CS₁₀ for that bit.

FIG. 12 shows an example of an analog-to-digital conversion methodincluding a digital-to-analog conversion method. Although the followingoperations are primarily described with respect to the implementationsof FIGS. 9 and 11, the operations are easily modified to apply to otherimplementations of the present disclosure. In one embodiment, theoperations are iteratively performed. The method begins at 450. At 452,the S/H circuit 302 receives the analog signal LANEINP, LANEINN, whichis the input differential analog signal having positive and negativepotentials, as shown in FIG. 11.

In one embodiment, the following operations 456, 458, 460, 462 areperformed during the same (first) period of time. At 456, the S/Hcircuit 302 including switches 370, 372 samples and holds a voltage ofthe analog signal LANEINP, LANEINN. At 458, capacitors CS₁₀ of theCS-DAC 362 are charged based on reference voltages V_(REFP), V_(REFN).At 460, capacitors CR₆ of the CR-DAC 364 are charged based on the commonmode voltage VCM. At 462, capacitors CΔΣ of the ΔΣ-DAC 366 are chargedbased on the common mode voltage VCM.

At 464, the hybrid DAC 314 or 354 performs a conversion for current bitsduring a second period of time. If a MSB of a predetermined number ofMSBs is being converted, then the SAR-ΔΣ control module 360 at 464Agenerates the bit control signals b[1:n] such that charges on thecapacitors CS₁₀ of CS-DAC 362 are shared by changing states of switches380, 382. If intermediary bits or one or more LSBs are being converted,then the SAR-A control module 312, 360 at 464B generates the bit controlsignals b[1:n] such that charges on the capacitors CR₆ are redistributedby changing states of the switches 402, 404 to connect the capacitorsCR₆ to received reference voltages V_(REFN), V_(REFP). If a LSB of apredetermined number of LSBs is being converted, then the SAR-A controlmodule 360 at 464C generates the bit control signals b[1:n] such thatcharges on the capacitors CΔΣ are redistributed by changing states ofthe switches 432, 434 to connect the capacitors CΔΣ to receivedreference voltages V_(REFN), V_(REFP).

At 466, output of the hybrid DAC 314, 354 is provided to the amplifier306, 356. The output is residual voltage Vr (or VRP, VRN). At 468, thelatch 308, 358 latches the residual voltage Vr based on the clock signalClk. At 470, the SAR-A control module 312, 360 performs a successiveapproximation algorithm or ΔΣ algorithm based on the clock signal Clkand latched amplified output of the latch 308, 358. The successiveapproximation algorithm includes, for example, converting an analogsignal into a discrete digital representation via a binary search of allpossible quantization levels before finally converging upon a digitaloutput for each conversion.

At 472, the SAR-ΔΣ control module 312, 360 determines whether anothercycle for a current conversion is to be performed. If another cycle fora current conversion is to be performed, task 464 is performed,otherwise task 473 is performed. At 473, the SAR-ΔΣ control module 312,360 outputs a digital signal representing a word converted by theSAR-ADC 300, 351.

At 474, the SAR-ΔΣ control module 312, 360 determines whether anotherconversion is to be performed. If conversion is to be performed, task476 is performed, otherwise the method ends at 480. At 476, a conversioncount is incremented by the SAR-ΔΣ control module 312, 360.

The above-described operations are meant to be illustrative examples; inone or more embodiments, the operations are performed sequentially,synchronously, simultaneously, continuously, during overlapping timeperiods or in a different order depending upon the application. Also, inone embodiment, one or more of the operations is not performed orskipped depending on the implementation and/or sequence of events.

The below described examples include a 12-bit 4-way interleaved ADC,which is shown in FIG. 13. The ADC performs, as an example, 600mega-samples per second (MS/s) conversions. Although a 12-bit 4-way(i.e. 4 lanes) interleaved ADC is described, the aspects of the presentdisclosure are applicable to ADCs converting a different number of bitsin parallel and having a different number of lanes.

FIG. 13 shows SAR-ΔΣ ADC 500 that includes a clock generator 502, aninput buffer 504, a reference buffer 506, Lanes 1-4, a serializer andgain correction module 508 and a lane gain calibration module 510. Theclock generator 502 receives a clock signal Clk at a resonant frequencyfs and generates phase shifted clock signals for the lanes 1-4. Thephase shifted clock signals have phases 0°, 90°, 180° and 270°, whichare provided respectively to the lanes 1-4. The input buffer 504receives an analog input signal ADC IN and provides the analog inputsignal ADC IN to the lanes 1-4. The reference buffer 506 receives abandgap signal BG and provides the bandgap signal BG to the lanes 1-4.An output of the reference buffer is provided across a referencecapacitor CREF.

Each of the lanes 1-4 includes a S/H circuit 512, a first multiplier514, a first subtractor 516, an integrator 518, a second subtractor 520,a latch 522, a latch offset calibration module 524, a SAR-ΔΣ controlmodule 526, and a SAR-ΔΣ DAC 528. The S/H circuit 512 samples the analoginput voltage ADC IN. The multiplier 514 multiplies an output of the S/Hcircuit 512 and a pseudo-random bit sequence (PRBS) generated by a PRBSgenerator 530. The first subtractor 516 subtracts an output of theSAR-ΔΣ DAC 528 from an output of the multiplier 516. The integrator 518integrates an output of the first subtractor 516. The second subtractor520 subtracts an output of the latch offset calibration module 524 froman output of the integrator 518. The latch 522 latches an output of thesecond subtractor 520 and provides the latched output to the SAR-ΔΣcontrol module 526. The SAR-A control module 526 and the SAR-ΔΣ DAC 528are configured and/or operate similarly to (i) the SAR-ΔΣ control module312 and the SAR-ΔΣ DAC 314 of FIG. 9, and/or (ii) the SAR-ΔΣ controlmodule 360 and the hybrid DAC 354 of FIG. 11.

Each of the lanes 1-4 further include ΔΣ decimation module 530, a SARweighted summation module 532, a SAR-SD DAC calibration module 534, asummer 536, a lane offset calibration module 538, and a secondmultiplier 540. The ΔΣ decimation module 530 filters a bit signal bsdout of the SAR-ΔΣ control module 526. The SAR weighted summation module532 provides a weighted sum of the bit signals bsar received from theSAR-ΔΣ control module 360. The SAR-SD DAC calibration module 534generates SAR weights, which are provided to the SAR weighted summationmodule 532, which performs the weighted summation based on the SARweights. The summer 536 sums outputs of the ΔΣ decimation module 530 andthe SAR weighted summation module 532.

The lane offset calibration module 538 includes an average module 542and a third subtractor 544. The average module 542 averages outputs ofthe summer 536. The third subtractor 544 subtracts the average output ofthe average module 542 from the summation output of the summer 536. Thesecond multiplier 540 multiples an output of the third subtractor 544 bythe PRBS to provide a digital output. Digital outputs of the lanes 1-4are provided to the serializer and gain correction module 508. The lanegain calibration module 510 generates a selection signal. The serializerand gain correction module 508 selects output of one of the lanes 1-4based on the selection signal and adjusts gain of the selected output toprovide an ADC output signal ADC OUT. In the example shown, the ADCoutput signal ADC OUT is a 12-bit digital signal. The serializer andgain correction module 508 serializes the parallel outputs of the lanes1-4 to provide a serial signal (i.e. the ADC output signal ADC OUT).

The ADC architecture of FIG. 13 includes 4-way interleaved ADC lanes(i.e. lanes 1-4) driven at 25% duty-cycle for maximized input signalbuffer loading. For each of the lanes 1-4 at start-up, an offset of thelatch 522 is calibrated by the latch offset calibration module 524 toprevent incremental saturation of the SAR-ΔΣ ADC 500. Next thecapacitive SAR-DAC is calibrated for linearity requirements. Finally,gain and offset mismatches among the lanes 1-4 are measured andcorrection coefficients are applied during operation. This is done bythe lane offset calibration module 538. Slow offset mismatch variationsare tracked and corrected.

FIG. 14 shows an example timing diagram of the signals φ_(TH), enSAR,enΔΣ. FIG. 14 further shows an example of percentages of time associatedwith S/H operation versus SAR-ΔΣ DAC operation for each of the lanes 1-4of FIG. 13. Offset operation of the lanes is also shown, which is due tothe S/H timing associated with each bit of the lanes 1-4.

In this application and in one or more embodiments, including thedefinitions below, the term “module” or the term “controller” isreplaced with the term “circuit.” The term “module” refers to, be partof, or include: an Application Specific Integrated Circuit (ASIC); adigital, analog, or mixed analog/digital discrete circuit; a digital,analog, or mixed analog/digital integrated circuit; a combinationallogic circuit; a field programmable gate array (FPGA); a processorcircuit (shared, dedicated, or group) that executes code; a memorycircuit (shared, dedicated, or group) that stores code executed by theprocessor circuit; other suitable hardware components that provide thedescribed functionality; or a combination of some or all of the above,such as in a system-on-chip.

What is claimed is:
 1. An analog-to-digital converter comprising: afirst analog-to-digital converter configured to receive an analog inputsignal and convert the analog input signal to a first digital signal,the first analog-to-digital converter comprising a successiveapproximation module, wherein the successive approximation module isconfigured to perform a successive approximation to generate the firstdigital signal; a second analog-to-digital converter configured toconvert an analog output of the first-analog-to-digital converter to asecond digital signal, wherein the analog output of the firstanalog-to-digital converter is generated based on the analog inputsignal, wherein the second analog-to-digital converter is a fineconversion analog-to-digital converter relative to the firstanalog-to-digital converter, wherein the second analog-to-digitalconverter performs the delta-sigma conversion process and comprises adecimation filter, and wherein the decimation filter is configured tosuppress noise which reduces amplification and power consumptionrequirements of the first analog-to-digital converter, and perform adelta-sigma decimation process to generate the second digital signalbased on the analog output of the first analog-to-digital converter; anda combination module configured to combine the first digital signal andthe second digital signal to provide a resultant output signal.
 2. Theanalog-to-digital converter of claim 1, wherein the secondanalog-to-digital converter comprises: a digital-to-analog converter, asubtractor configured to receive the output of the firstanalog-to-digital converter and an output of the digital-to-analogconverter, a comparator configured to receive an output of thesubtractor and a first reference voltage, wherein the comparatorcomprises a loop filter configured to filter the output of thesubtractor, and a latch configured to receive an output of the loopfilter, and wherein the decimation filter is configured to filter andperform the delta-sigma decimation process to generate the seconddigital signal based on an output of the latch, wherein the output ofthe second digital-to-analog converter is generated based on the seconddigital signal.
 3. The analog-to-digital converter of claim 1, whereinthe first analog-to-digital converter comprises: a firstdigital-to-analog converter, a first subtractor configured to receive asample and hold signal and an output of the first digital-to-analogconverter, and a first comparator configured to compare an output of thefirst subtractor with a first reference voltage, wherein the successiveapproximation module is configured to perform the successiveapproximation to generate the first digital signal based on an output ofthe first comparator, wherein the output of the first digital-to-analogconverter is generated based on the first digital signal.
 4. Theanalog-to-digital converter of claim 3, wherein the firstdigital-to-analog converter comprises a charge sharing digital-to-analogconverter or a charge redistribution digital-to-analog converterconfiguration.
 5. The analog-to-digital converter of claim 3, whereinthe second analog-to-digital converter comprises: a seconddigital-to-analog converter, a second subtractor configured to receivethe output of the first subtractor and an output of the seconddigital-to-analog converter, a second comparator configured to receivean output of the second subtractor and the first reference voltage,wherein the second comparator comprises a loop filter configured tofilter the output of the subtractor, and a latch configured to receivean output of the loop filter, and wherein the decimation filter isconfigured to perform the delta-sigma decimation process to generate thesecond digital signal based on an output of the latch, wherein theoutput of the second digital-to-analog converter is generated based onthe second digital signal.
 6. The analog-to-digital converter of claim5, wherein: the first analog-to-digital converter is configured to, fora plurality of cycles, generate the first digital signal based on thesample and hold signal; and the second analog-to-digital converter isconfigured to generate the second digital signal based on a differencebetween (i) the sample and hold signal and (ii) a result of a successiveapproximation performed subsequent to the plurality of cycles, whereinthe second analog-to-digital converter does not generate the seconddigital signal during the plurality of cycles.
 7. The analog-to-digitalconverter of claim 5, wherein: the first digital-to-analog converter isconfigured to convert a first most-significant-bit of a plurality ofbits to be converted to an analog output signal; the firstdigital-to-analog converter comprises a first plurality of capacitors;the first plurality of capacitors of the a first digital-to-analogconverter are charged by a plurality of reference voltages during asampling phase of the analog signal; the plurality of reference voltagesincludes the first reference voltage; charges of the first plurality ofcapacitors of the first digital-to-analog converter are shared duringsuccessive approximations of a first one or more bits of the firstdigital signal; the second digital-to-analog converter is configured toconvert a first least-significant-bit of the plurality of bits to beconverted to the analog output signal; the second digital-to-analogconverter comprises a second plurality of capacitors; the secondplurality of capacitors of the second digital-to-analog converter arecharged based on a common mode voltage during the sampling phase of theanalog output signal; and the second digital-to-analog converter isconfigured to perform charge redistribution by connecting the secondplurality of capacitors of the second digital-to-analog converter toreceive the plurality of reference voltages during successiveapproximations of a second one or more bits of the first digital signal.8. A method comprising: receiving an analog input signal and convertingthe analog input signal to a first digital signal at a firstanalog-to-digital converter; performing a successive approximation togenerate the first digital signal via the first analog-to-digitalconverter; converting an analog output of the first analog-to-digitalconverter to a second digital signal via a second analog-to-digitalconverter, wherein the second analog-to-digital converter is a fineconversion analog-to-digital converter relative to the firstanalog-to-digital converter; suppressing noise of the secondanalog-to-digital converter via a decimation filter; performing adelta-sigma conversion via the second analog-to-digital converter togenerate the second digital signal based on the analog output of thefirst analog-to-digital converter, wherein the analog output of thefirst digital-to-analog converter is generated based on the analog inputsignal; and combining the first digital signal and the second digitalsignal to provide a resultant output signal.
 9. The method of claim 8,further comprising: receiving a sample and hold signal and an output ofa first digital-to-analog converter at a first subtractor; comparing anoutput of the first subtractor with a first reference voltage at a firstcomparator, wherein the first analog-to-digital converter comprises thefirst subtractor, the first comparator and the first analog-to-digitalconverter; and performing the successive approximation to generate thefirst digital signal based on an output of the first comparator, whereinthe output of the first digital-to-analog converter is generated basedon the first digital signal.
 10. The method of claim 9, furthercomprising: receiving the output of the first subtractor and an outputof a second digital-to-analog converter at a second subtractor;receiving an output of the second subtractor and the first referencevoltage at a second comparator; filter the output of the subtractor viaa loop filter of the second comparator; latching an output of the loopfilter via a latch of the second comparator; and performing thedelta-sigma conversion process to generate the second digital signalbased on an output of the latch, wherein the output of the seconddigital-to-analog converter is generated based on the second digitalsignal.
 11. The method of claim 8, further comprising: converting afirst most-significant-bit of a plurality of bits to be converted to ananalog output signal via the first digital-to-analog converter; charginga first plurality of capacitors of the first digital-to-analog converterby a plurality of reference voltages during a sampling phase of theanalog signal; sharing charges of the first plurality of capacitors ofthe first digital-to-analog converter during successive approximationsof a first one or more bits of the first digital signal; converting afirst least-significant-bit of the plurality of bits to be converted tothe analog output signal via the second digital-to-analog converter;charging second plurality of capacitors of the second digital-to-analogconverter based on a common mode voltage during the sampling phase ofthe analog output signal; and performing charge redistribution byconnecting the second plurality of capacitors of the seconddigital-to-analog converter to receive the plurality of referencevoltages during successive approximations of a second one or more bitsof the first digital signal.
 12. An analog-to-digital convertercomprising: a digital-to-analog converter circuit configured to converta plurality of bits of a digital signal to an analog signal; a sampleand hold circuit to sample an analog input signal; a subtractorconfigured to subtract the analog signal from an output of the sampleand hold circuit; wherein the digital-to-analog converter circuitcomprises a first digital-to-analog converter configured to convert afirst most-significant-bit of the plurality of bits; a seconddigital-to-analog converter configured to convert a firstleast-significant-bit of the plurality of bits, wherein the seconddigital-to-analog converter is a delta-sigma digital-to-analogconverter; an amplifier configured to amplify an output of thedigital-to-analog converter circuit; a latch configured to latch anoutput of the amplifier; and a successive approximation moduleconfigured to (i) receive an output of the latch, and (ii) performsuccessive approximations to generate the digital signal.
 13. Theanalog-to-digital converter circuit of claim 12, wherein the amplifieris configured to amplify an output of the first digital-to-analogconverter and an output of the second digital-to-analog converter. 14.The analog-to-digital converter circuit of claim 12, wherein the firstdigital-to-analog converter is a charge sharing digital-to-analogconverter.
 15. The analog-to-digital converter circuit of claim 12,wherein: the first digital-to-analog converter comprises a plurality ofcircuits corresponding to some of the plurality of bits; each of theplurality of circuits is configured to convert a respective one of aplurality of most-significant-bits of the plurality of bits; theplurality of most-significant-bits comprise the firstmost-significant-bit; the second digital-to-analog converter comprisesone or more circuits corresponding to one or more of the plurality ofbits; each of the one or more circuits is configured to convert arespective one of a plurality of least-significant-bits of the pluralityof bits; the plurality of least-significant-bits comprise the firstleast-significant-bit; and the plurality of circuits do not include theone or more circuits.
 16. The analog-to-digital converter circuit ofclaim 12, further comprising a capacitor, a switch and a NOR gate,wherein: the amplifier comprises an integrator; the capacitor isconnected between an output of the integrator and a reference terminal;and the switch is connected across the capacitor and receives a controlsignal from the NOR gate.
 17. The analog-to-digital converter circuit ofclaim 16, further comprising a control module, wherein: the firstdigital-to-analog converter comprises a first plurality of switches; thesecond digital-to-analog converter comprises a second plurality ofswitches; and the control module is configured to, based on the outputof the latch, control states of the first plurality of switches andstates of the second plurality of switches.
 18. The analog-to-digitalconverter circuit of claim 16, further comprising a control module whichsets an input of the NOR gate based on the output of the latch.
 19. Theanalog-to-digital converter circuit of claim 16, further comprising acontrol module: the control module comprises a decimation filter; andthe decimation filter is configured to suppress noise which reducesamplification and power consumption requirements of the firstanalog-to-digital converter.